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mic24051 12v, 6 a high - efficiency buck regulator superswitcher ii ? hyperlight load is a registered tr ademark of micrel, inc . hyper speed control, superswitcher ii, and any capacitor are trademarks of micrel, inc. micrel inc. ? 2180 fortune drive ? san jose, ca 95131 ? usa ? tel +1 (408) 944 - 0800 ? fax + 1 (408) 474 - 1000 ? http://www.micrel.com november 2012 m9999 -1 12612 -a general description the micrel mic24051 is a constant - frequency, synchronous buck regulator featuring a unique adaptive on - time control architecture . the mic24051 operates over an input supply range of 4.5v to 19v and provides a regulated output of up to 6 a of output current. the output voltage is adjustable down to 0.8v with a guaranteed accuracy of 1%, and the device operates at a switching frequency of 6 00khz. micrel?s hyper speed control ? architecture allows for ultra - fast transient response while reducing the output capacitance and also makes ( high v in )/( low v out ) operation possible. this adaptive t on ripple control architecture combines the advantages of fixed - frequency operation and fast transient response in a single device . the mic24051 offers a full suite of protection features to ensure protection of the ic during fault conditions. these include undervoltage lockout to ensure proper operation und er power - sag conditions, internal soft - start to reduce inrush current, fold b ack current limit , ?hicc up mode ? short - circuit protection and thermal shutdown. an open - drain power good (pg) pin is provided. the 6a hyperlight load ? part, mic24052, is also available on micrel?s web site. all support documentation can be found on micrel? s web site at: www.micrel.com . features ? hyper speed control architecture enables - high d elta v operation (v in = 19v and v out = 0.8v) - small output capacitance ? 4.5v to 19v voltage input ? 6a output current capability, up to 95% efficiency ? adjustable output from 0.8v to 5.5v ? 1% feedback accuracy ? any capacitor ? s table - z ero - to - high esr ? 600khz switching frequency ? no ex ternal compe nsation ? power good (pg) output ? fold back current - limit and ?hiccup mode ? short - circuit protectio n ? supports safe start - up into a pre - biased load ? ? 40 c to +125 c junction temperature range ? available in 28 - pin 5mm 6mm qfn package applications ? servers and work stations ? routers, switches, and t elecom equipment ? base station s _________________________ ___________________________________________________ ________________ _____________________________ typical application efficiency (vin = 12v) vs. output current 50 55 60 65 70 75 80 85 90 95 100 0 1 2 3 4 5 6 7 8 output current (a) efficiency (%) 5.0v 3.3v 2.5v 1.8v 1.5v 1.2v 1.0v 0.9v 0.8v
micrel, inc. mic2 4051 november 2012 2 m9999 -1 12612 -a ordering information part number switching frequency voltage package junction temperature range lead finish mic24 05 1yjl 600k hz adjustable 28- pin 5mm 6mm qfn ? 40 c to + 125 c pb - free pin configuration 28- pin 5mm 6mm qfn ( jl) (top view) pin description pin number pin name pin function 1 pvdd 5v internal linear regulator o utput . pvdd supply is the power mosfet gate d rive supply voltage and created by internal ldo from vin. when vin < +5.5v, pvdd should be tied to pvin pins. a 2.2f ceramic capacitor from the pvdd pin to pgnd (pin 2) must be place next to the ic. 2, 5, 6, 7, 8, 21 pgnd power ground. pgnd is the groun d path for the mic24 05 1 buck converter power stage. the pgnd pins connect to the low - side n - channel internal mosfet gate drive supply ground, the sources of the mosfets, the negative terminals of input capacitors, and the negative t erminals of output capa citors. the loop for the power ground should be as small as possible and separate from the signal ground (sgnd) loop. 3 nc no connect. 4, 9, 10, 11, 12 sw switch node o utput . internal connection for the high - side mosfet source and low - side mosfet drai n. due to the high - speed switching on this pin, the sw pin should be routed away from sensitive nodes. 13,14,15, 16,17,18,19 pvin high - side n - int ernal mosfet drain connection input. the pv in operating voltage range is from 4.5v to 19v. input capacit ors between the pv in pins and the power ground (pgnd) are required and keep the connection short. 20 bst boost o utput . bootstrapped voltage to the high - side n - channel mosfet driver. a schottky diode is connected between the pvdd pin and the bst pin. a b oost capacitor of 0.1f is connected between the bst pin and the sw pin. adding a small resistor at the bst pin can slow down the turn - on time of high - side n - channel mosfets.
micrel, inc. mic2 4051 november 2012 3 m9999 -1 12612 -a pin description (continued) pin number pin name pin function 22 cs current sen se i nput . the cs pin senses current by monitoring the voltage across the low - side mosfet during the off - time. the current sensing is necessary for short circuit protection. in order to sense the current accurately, connect the low - side mosfet drain to sw using a kelvin connection. the cs pin is also the high - side mosfet?s output driver return. 23 sgnd signal g round. sgnd must be connected directly to the ground planes. do not route the sgnd pin to the pgnd pad on the top layer (see pcb layout guidelines for details). 24 fb feedback i nput . input to the transconductance amplifier of the control loop. the fb pin is regulated to 0.8v. a resistor divider connecting the feedback to the output is used to adjust the desired output voltage. 25 pg power good o utput . open drain o utput. the pg pin is externally tied with a resistor to vdd. a high output is asserted when v out > 92% of nominal. 26 en enable input . a logic level control of the output. the en pin is cmos - compatible. logic high = enable, logic low = shutdown. in the off state, supply current of the device is greatly reduced (typically 5a). the en pin should not be left floating . 27 vin power supply voltage i nput . requires bypass capacitor to sgnd. 28 vdd 5v internal linear regulator o utput . vdd supply is the power mosfet gate drive supply voltage and the supply bus for the ic . vdd is created by internal ldo from vin . when v in < +5.5v , vdd s hould be tied to pvin pins. a 1 f ceramic capacitor from the vdd pin to s gnd pins must be place next to the ic.
micrel, inc. mic2 4051 november 2012 4 m9999 -1 12612 -a absolute maximum ratings (1) pvin to pgnd ............................................... ? 0.3v to +2 9 v v in to pgnd ................................................. ? 0.3v to pv in p v dd , v dd to pgnd ..................................... ? 0.3v to +6v v sw , v cs to pgnd ............................. ? 0.3v to (pv in +0.3v) v bst to v sw ........................................................ ? 0.3v to 6v v bst to pgnd .................................................. ? 0.3v to 3 5 v v fb , v pg to pgnd ............................. ? 0.3v to (v dd + 0.3v) v en to pgnd ....................................... ? 0.3v to (v in +0.3v) pgnd to sgnd ............................................ ? 0.3v to +0.3v junction temperature .............................................. +150c storage temperature (t s ) ......................... ? 65 c to +150 c le ad temperature (soldering, 10s ) ............................ 260c esd rating ( 2) ?? ?????????? esd sensitive operating ratings (3) supply voltage ( pv in , v in ) ........................... 4.5v to 19v pvdd, vdd supply voltage ( p v dd, v dd) .. 4.5v to 5.5v enable input (v en ) .............................................. 0v to v in junction temperature (t j ) ..................... ? 40 c to +125 c maximum power dissipation .................................. note 4 package thermal resistance (4) 5mm x 6mm qfn - 28 ( ja ) ............................. 28 c/w 5mm x 6mm qfn - 28 ( jc ) ........................... 2.5 c/w electrical characteristics ( 5 ) pvin = v in = v en = 12v, v bst ? v sw = 5v; t a = 25 c, unless noted. bold values indicate ? 40c t j +125 c. parameter condition min . typ . max . units power supply input input voltage range (v in , pv in ) 4.5 19 v quiescent supply current v fb = 1.5v (non - switching) 730 1500 a shutdown supply current v en = 0v 5 10 a vdd supply voltage vdd output voltage vin = 7v to 19v , i dd = 40ma 4. 8 5 5.4 v vdd uvlo threshold vdd rising 3.7 4.2 4.5 v vdd u vlo hysteresis 400 mv dropout voltage (v in ? vdd) i dd = 25ma 380 600 mv dc/dc controller output - voltage adjust range (v out ) 0.8 5.5 v reference feedback reference voltage 0 c t j 85c ( 1.0%) 0.792 0.8 0.808 v ? 40c t j 125 c ( 1.5%) 0.788 0.8 0.812 load regulation i out = 0 a to 6a ( continuous mode ) 0.25 % line regulation vin = 4.5v to 19v 0.25 % fb bias current v fb = 0.8v 50 500 na notes: 1. exceeding the absolute maximum rating may damage the device. 2 . devic es are esd sensitive. handling precautions recommended. human body model, 1.5k ? in series with 100pf. 3. the device is not guaranteed to function outside operating range. 4. pd (max) = (t j (max) ? t a )/ ja , where ja depends upon the printed circuit layout. a 5 square inch 4 layer, 0.62?, fr - 4 pcb with 2oz finish copper weight per layer is used for the ja . 5 . specification for packaged product only.
micrel, inc. mic2 4051 november 2012 5 m9999 -1 12612 -a electrical characteristics (5 ) (continued) pvin = v in = v en = 12v, v bst ? v sw = 5v; t a = 25 c, unless noted. bold values indicate ? 40c t j +125 c. parameter condition min. typ. max. units enable control en logic level high 1.8 v en logic level low 0.6 v en bias current v en = 12v 6 30 a oscillator switching frequency (6 ) v out = 2.5v 450 600 750 kh z maximum duty cycle (7) v fb = 0v 82 % minimum duty cycle v fb = 1.0v 0 % minimum off - time 300 ns soft - start soft - start time 3 ms short - circuit protection peak inductor current - limit threshold v fb = 0.8v, t j = 25 c 7.5 11 17 a v fb = 0.8 v, t j = 125c 6.6 11 17 short - circuit current v fb = 0v 8 a internal fets top - mosfet r ds (on) i sw = 1a 42 m ? bottom - mosfet r ds (on) i sw = 1a 12.5 m ? sw leakage current v en = 0v 60 a v in leakage current v en = 0v 25 a power good (pg) pg threshold voltage sweep v fb from low to high 85 92 95 %v out pg hysteresis sweep v fb from high to low 5.5 %v out pg delay time sweep v fb from low to high 100 s pg low voltage sweep v fb < 0.9 v nom , i pg = 1ma 70 200 mv thermal protection over - temperature shutdown t j rising 160 c over - temperature shutdown hysteresis 15 c notes: 6. measured in test mod e. 7. the maximum duty - cycle is limited by the fixed mandatory off - time t off of typically 300ns.
micrel, inc. mic2 4051 november 2012 6 m9999 -1 12612 -a typical characteristics vin operating supply current vs. input voltage 0 4 8 12 16 20 4 7 10 13 16 19 input voltage (v) supply current (ma) v out = 1.8v i out = 0a switching vin shutdown current vs. input voltage 0 15 30 45 60 4 7 10 13 16 19 input voltage (v) shutdown current (a) v en = 0v r en = open vdd output voltage vs. input voltage 0 2 4 6 8 10 4 7 10 13 16 19 input voltage (v) vdd voltage (v) v fb = 0.9v i dd = 10ma feedback voltage vs. input voltage 0.792 0.796 0.800 0.804 0.808 4 7 10 13 16 19 input voltage (v) feedback voltage (v) v out = 1.8v i out = 0a total regulation vs. input voltage -1.0% -0.5% 0.0% 0.5% 1.0% 4 7 10 13 16 19 input voltage (v) total regulation (%) v out = 1.8v i out = 0a to 6a output current limit vs. input voltage 0 5 10 15 20 4 7 10 13 16 19 input voltage (v) current limit (a) v out = 1.8v switching frequency vs. input voltage 350 400 450 500 550 600 650 700 750 4 7 10 13 16 19 input voltage (v) frequency (khz) v out = 1.8v i out = 0a enable input current vs. input voltage 0 4 8 12 16 4 7 10 13 16 19 input voltage (v) en input current (a) v en = vin 80% 85% 90% 95% 100% 4 7 10 13 16 19 v pg threshold/v ref (%) input voltage (v) pg/v ref ratio vs. input voltage v fb = 0.8v
micrel, inc. mic2 4051 november 2012 7 m9999 -1 12612 -a typical characteristics (continued) vin operating supply current vs. temperature 0 4 8 12 16 20 -50 -25 0 25 50 75 100 125 temperature (c) supply current (ma) vin = 12v v out = 1.8v i out = 0a switching 0 5 10 15 20 -50 -25 0 25 50 75 100 125 supply current ( a) temperature ( c) vin shutdown current vs. temperature vin = 12v i out = 0a v en = 0v vdd uvlo threshold vs. temperature 0 1 2 3 4 5 -50 -25 0 25 50 75 100 125 temperature (c) vdd threshold (v) rising falling hyst feedback voltage vs. temperature 0.792 0.796 0.800 0.804 0.808 -50 -25 0 25 50 75 100 125 temperature (c) feeback voltage (v) vin = 12v v out = 1.8v i out = 0a load regulation vs. temperature -1.0% -0.5% 0.0% 0.5% 1.0% -50 -25 0 25 50 75 100 125 temperature (c) load regulation (%) vin = 12v v out = 1.8v i out = 0a to 6a line regulation vs. temperature -0.2% -0.1% 0.0% 0.1% 0.2% 0.3% 0.4% -50 -25 0 25 50 75 100 125 temperature (c) line regulation (%) vin = 4.5v to 19v v out = 1.8v i out = 0a switching frequency vs. temperature 500 550 600 650 700 -50 -25 0 25 50 75 100 125 temperature (c) frequency (khz) vin = 12v v out = 1.8v i out = 0a vdd vs. temperature 2 3 4 5 6 -50 -25 0 25 50 75 100 125 temperature (c) vdd (v) vin = 12v i out = 0a output current limit vs. temperature 0 5 10 15 20 25 -50 -25 0 25 50 75 100 125 temperature (c) current limit (a) vin = 12v v out = 1.8v
micrel, inc. mic2 4051 november 2012 8 m9999 -1 12612 -a typ ical characteristics (continued) switching frequency vs.output voltage 100 200 300 400 500 600 700 0.0 0.5 1.0 1.5 2.0 2.5 3.0 3.5 4.0 4.5 5.0 output voltage (v) frequency (khz) vin = 12 i out = 0a feedback voltage vs. output current 0.792 0.796 0.800 0.804 0.808 0 1 2 3 4 5 6 output current (a) feedback voltage (v) vin = 12v v out = 1.8v output voltage vs. output current 1.782 1.786 1.790 1.794 1.798 1.802 1.806 1.810 1.814 1.818 0 1 2 3 4 5 6 output current (a) output voltage (v) vin = 12v v out = 1.8v line regulation vs. output current -1.0% -0.5% 0.0% 0.5% 1.0% 0 1 2 3 4 5 6 output current (a) line regulation (%) vin = 4.5v to 19v v out = 1.8v switching frequency vs. output current 500 550 600 650 700 0 1 2 3 4 5 6 output current (a) frequency (khz) vin = 12v v out = 1.8v efficiency (vin = 5v) vs. output current 50 55 60 65 70 75 80 85 90 95 100 0 1 2 3 4 5 6 7 8 output current (a) efficiency (%) 3.3v 2.5v 1.8v 1.5v 1.2v 1.0v 0.9v 0.8v ic power dissipation (vin = 5v) vs. output current 0.0 0.5 1.0 1.5 2.0 2.5 0 1 2 3 4 5 6 output current (a) ic power dissipation (w) vin = 5v v out = 3.3v v out = 0.8v die temperature* (vin = 5v) vs. output current 0 10 20 30 40 50 60 0 1 2 3 4 5 6 output current (a) die temperature (c) vin = 5v v out = 1.8v die temperature* : the temperature measurement was taken at the hottest point on the mic24051 case mounted on a 5 square inch 4 layer, 0.62?, fr - 4 pcb with 2oz finish copper weight per layer, see thermal measurement section. actual results will depend upon the size of the pcb, amb ient temperature and proximity to other heat emitting components. 3 3.4 3.8 4.2 4.6 5 0 1 2 3 4 5 6 7 8 output voltage (v) output current (a) output voltage (vin = 5v) vs. output current t a 25oc 85oc 125oc vin = 5v v fb < 0.8v
micrel, inc. mic2 4051 november 2012 9 m9999 -1 12612 -a typical characteristics (continued) efficiency (vin = 12v) vs. output current 50 55 60 65 70 75 80 85 90 95 100 0 1 2 3 4 5 6 7 8 output current (a) efficiency (%) 5.0v 3.3v 2.5v 1.8v 1.5v 1.2v 1.0v 0.9v 0.8v ic power dissipation (vin = 12v) vs. output current 0.0 0.5 1.0 1.5 2.0 2.5 0 1 2 3 4 5 6 output current (a) ic power dissipation (w) vin = 12v v out = 5.0v v out = 0.8v die temperature* (vin = 12v) vs. output current 0 10 20 30 40 50 60 0 1 2 3 4 5 6 output current (a) die temperature (c) vin = 12v v out = 1.8v thermal derating* vs. ambient temperature 0 2 4 6 8 10 12 -50 -25 0 25 50 75 100 125 ambient temperature (c) output current (a) 1.5v v in = 5v v out = 0.8, 1.2, 1.5v 0.8v thermal derating* vs. ambient temperature 0 2 4 6 8 10 12 -50 -25 0 25 50 75 100 125 ambient temperature (c) output current (a) v in = 5v v out = 1.8, 2.5, 3.3v 3.3v 1.8v thermal derating* vs. ambient temperature 0 2 4 6 8 10 12 -50 -25 0 25 50 75 100 125 ambient temperature (c) output current (a) 1.8v 0.8v v in = 12v v out = 0.8, 1.2, 1.8v thermal derating* vs. ambient temperature 0 2 4 6 8 10 12 -50 -25 0 25 50 75 100 125 ambient temperature (c) output current (a) 5v 2.5v v in = 12v v out = 2.5, 3.3, 5v die temperature* : the temperature measurement was taken at the hottest point on the mic24051 case mounted on a 5 square inch 4 layer, 0.62?, fr - 4 pcb with 2oz finish copper weight per layer, see thermal measurement section. act ual results will depend upon the size of the pcb, ambient temperature and proximity to other heat emitting components.
micrel, inc. mic2 4051 november 2012 10 m9999 -1 12612 -a functional characteristics
micrel, inc. mic2 4051 november 2012 11 m9999 -1 12612 -a functional characteristics (continued)
micrel, inc. mic2 4051 november 2012 12 m9999 -1 12612 -a functional characteristics (continued)
micrel, inc. mic2 4051 november 2012 13 m9999 -1 12612 -a functional diagram figure 1. mic24051 block diagram
micrel, inc. mic2 4051 november 2012 14 m9999 -1 12612 -a functional description the mic24051 is an adaptive on - time synchronous step - down dc / dc regulator with an internal 5v linear regulator and a powe r good (pg) output. it is designed to operate over a wide input voltage range from 4.5v to 19v and provides a regulated output voltage at up to 6 a of output current. a n adaptive on - time control scheme is employed in to obtain a constant switching frequency and to simplify the control compensation. overcurrent protection is implemented without the use of an external sense resistor. the device includes an internal soft - start function which reduces the power supply input surge current at start - up by controllin g the output voltage rise time. theory of operation the mic24051 operates in a continuous mode as shown in figure 1. continuous mode in continuous mode, the output voltage is sensed by the mic24051 feedback pin fb via the voltage divider r1 and r2, and compared to a 0.8v reference voltage v ref at the error comparator through a low gain transconductance (g m ) amplifier. if the feedback voltage decreases and the output of the g m amplifier is below 0.8v, then the error comparator will trigger th e control logic and generate an on - time period. the on - time period length is predetermined by the ?fixed t on estimation? circuitry: khz 600 v v t in out ) estimated ( on = eq. 1 where v out is the output voltage and v in is the power stage input voltage. at the end of th e on - time period, the internal high - side driver turns off the high - side mosfet and the low - side driver turns on the low - side mosfet. the off - time period length depends upon the feedback voltage in most cases. when the feedback voltage decreases and the out put of the g m amplifier is below 0.8v, the on - time period is triggered and the off - time period ends. if the off - time period determined by the feedback voltage is less than the minimum off - time t off(min) , which is about 300ns, the mic24051 control l ogic will apply the t off(min) instead. t off(min) is required to maintain enough energy in the boost capacitor (c bst ) to drive the high - side mosfet. the maximum du ty cycle is obtained from the 30 0ns t off(min) : s s ) min ( off s max t ns 300 1 t t t d ? = ? = eq. 2 where t s = 1/ 600khz = 1.66 s. it is not recommended to use mic24051 with a off - time close to t off(min) during steady - state operation. also, as v out increases, the internal ripple injection will increase and reduce the line regulation performance. therefore, the maximum ou tput voltage of the mic24051 should be limited to 5.5v and the maximum external ripple injection should be limited to 200mv. please refer to ?setting output voltage? subsection in application information for more details. the actual on - time and res ulting switching frequency will vary with the part - to - part variation in the rise and fall times of the internal mosfets, the output load current, and variations in the v dd voltage. also, the minimum t on results in a lower switching frequency in high v in to v out applications, such as 18 v to 1.0v. the minimum t on measured on the mic24051 evaluation board is about 100ns. during load transients, the switching frequency is changed due to the varying off - time. to illustrate the control loop operation, we will analyze both the steady - state and load transient scenarios. figure 2 shows the mic24051 control loop timing during steady - state operation. during steady - state, the g m amplifier senses the feedback voltage ripple, which is proportional to the output voltage ripple and the inductor current ripple, to trigger the on - time period. the on - time is predetermined by the t on estimator . the termination of the off - time is controlled by the feedback voltage. at the valley of the feedback voltage ripple, wh ich occurs when v fb falls below v ref , the off period ends and the next on - time period is triggered through the control logic circuitry.
micrel, inc. mic2 4051 november 2012 15 m9999 -1 12612 -a figure 2. mic24051 control loop timing figure 3 shows the operation of the mic24051 during a load transient. the output voltage drops due to the sudden load increase, which causes the v fb to be less than v ref . this will cause the error comparator to trigger an on - time period. at the end of the on - time period, a minimum off - time t off(min) is generated to charge c bst since the feedback voltage is still below v ref . then, the next on - time period is triggered due to the low feedback voltage. therefore, the switching frequency changes during the load transient, but returns to the nominal fixed frequency once the output has stabilized at the new load current level. with the varying duty cycle and switching frequency, the output recovery time is fast and the output voltage deviation is small in mic24051 converter. figure 3. mic24051 load tr ansient response unlike true current - mode control, the mic24051 uses the output voltage ripple to trigger an on - time period. the output voltage ripple is proportional to the inductor current ripple if the esr of the output capacitor is large enou gh. the mic24051 control loop has the advantage of eliminating the need for slope compensation. in order to meet the stability requirements, the mic24051 feedback voltage ripple should be in phase with the inductor current ripple and larg e enough to be sensed by the g m amplifier and the error comparator. the recommended feedback voltage ripple is 20mv~100mv. if a low - esr output capacitor is selected, then the feedback voltage ripple may be too small to be sensed by the g m amplifier and the error comparator. also, the output voltage ripple and the feedback voltage ripple are not necessarily in phase with the inductor current ripple if the esr of the output capacitor is very low. in these cases, ripple injection is required to ens ure proper o peration. please refer to ?ripple injection? subsection in application information for more details about the ripple injection technique. v dd regulator the mic24051 provides a 5v regulated output for input voltage v in ranging from 5.5v to 19v . wh en v in < 5.5v, v dd should be tied to pv in pins to bypass the internal linear regulator . soft - start soft - start reduces the power supply input surge current at startup by controlling the output voltage rise time. the input surge appears while the output cap acitor is charged up. a slower output rise time will draw a lower input surge current. the mic24051 implements an internal digital soft - start by making the 0.8v reference voltage v ref ramp from 0 to 100% in about 3ms with 9.7mv steps. therefore, t he output voltage is controlled to increase slowly by a stair - case v fb ramp. once the soft - start cycle ends, the related circuitry is disabled to reduce current consumption. v dd must be powered up at the same time or after v in to make the soft - start functi on correctly. current limit the mic24051 uses the r ds(on) of the internal low - side power mosfet to sense over - current conditions. this method will avoid adding cost, board space and power losses taken by a discrete current sense resistor. the low - s ide mosfet is used because it displays much lower parasitic oscillations during switching than the high - side mosfet. in each switching cycle of the mic24051 converter, the inductor current is sensed by monitoring the low - side mosfet in the off per iod. if the peak ind uctor current is greater than 1 1 a, then the mic24051 turns off the high - side mosfet and a soft - start sequence is triggered. this mode of operation is called ?hiccup mode? and its purpose is to protect the downstream load in case of a hard short. the load current - limit threshold has a fold - back characteristic re lated to the feedback voltage a s shown in figure 4.
micrel, inc. mic2 4051 november 2012 16 m9999 -1 12612 -a current limit thresold vs. feedback voltage 0.0 4.0 8.0 12.0 16.0 20.0 0.0 0.2 0.4 0.6 0.8 1.0 feedback voltage (v) current limit threshold (a) figure 4. mic24051 current - limit foldback characteristic power good (pg) the power good (pg) pin is an op en drain output which indicates logic high when the output is nominally 92% of its steady state voltage. a pull - up resistor of more than 10k should be connected from pg to vdd. mosfet gate drive the block diagram ( figure 1 ) shows a bootstrap circuit, consisting of d1 (a schottky diode is recommended) and c bst . this circuit supplies energy to the high - side drive circuit. capacitor c bst is cha rged, while the low - side mosfet is on, and the voltage on the sw pin is approximately 0v. when the high - side mosfet driver is turned on, energy from c bst is used to turn the mosfet on. as the high - side mosfet turns on, the voltage on the sw pin increases t o approximately v in . diode d1 is reverse biased and c bst floats high while continuing to keep the high - side mosfet on. the bias current of the high - side driver is less than 10ma so a 0.1f to 1f is sufficient to hold the gate voltage with minimal droop for the power stroke (high - side switching) cycle, i.e. bs t = 10ma x 1.67s/0.1f = 167mv. when the low - side mosfet is turned back on, c bst is recharged through d1. a small resistor r g , which is in series with c bst , can be used to slow down the turn - on time of the high - side n - channel mosfet. the drive voltage is derived from the v dd supply voltage. the nominal low - side gate drive voltage is v dd and the nominal high - side gate drive voltage is approximately v dd ? v diode , where v diode is the voltage drop across d1. an approximate 30ns delay between the h igh - side and low - side driver transitions is used to prevent current from simultaneously flowing unimpeded through both mosfets.
micrel, inc. mic2 4051 november 2012 17 m9999 -1 12612 -a application information inductor selection values for inductance, peak, and rms curr ents are required to select the output inductor. the input and output voltages and the inductance value determine the peak - to - peak inductor ripple current. generally, higher inductance values are used with higher input voltages. larger peak - to - peak ripple currents will increase the power dissipation in the inductor and mosfets. larger output ripple currents will also require more output capacitance to smooth out the larger ripple current. smaller peak - to - peak ripple currents require a larger inductance valu e and therefore a larger and more expensive inductor. a good compromise between size, loss and cost is to set the inductor ripple current to be equal to 20% of the maximum output current. the inductance value is calculated in equation 3. ) max ( out sw ) max ( in out ) max ( in out i % 20 f v ) v v ( v l ? = eq. 3 where: f sw = switching frequency, 600khz 20% = ratio of ac ripple current to dc output current v in(max) = maximum power stage input voltage the peak - to - peak inductor current ripple is: l f v ) v v ( v i sw ) max ( in out ) max ( in out ) pp ( l ? = ? eq. 4 the peak inductor curren t is equal to the average output current plus one half of the peak - to - peak inductor current ripple. ) pp ( l ) max ( out ) pk ( l i 5 . 0 i i ? + = eq. 5 the rms inductor current is used to calculate the i2r losses in the inductor. 12 i i i 2 ) pp ( l 2 ) max ( out ) rms ( l ? + = eq. 6 maximizing efficiency requires the proper selection of core material and minimizing the winding resistance. the high - frequency operation of the mic24051 requires the use of ferrite materials for all but the most cost sensitive applications. lower cost iron po wder cores may be used but the increase in core loss will reduce the efficiency of the power supply. this is especially noticeable at low output power. the winding resistance decreases efficiency at the higher output current levels. the winding resistance must be minimized although this usually comes at the expense of a larger inductor. the power dissipated in the inductor is equal to the sum of the core and copper losses. at higher output loads, the core losses are usually insignificant and can be ignored. at lower output currents, the core losses can be a significant contributor. core loss information is usually available from the magnetics vendor. copper loss in the inductor is calculated by equation 7: winding 2 ) rms ( l ) cu ( inductor r i p = eq. 7 the resistance of t he copper wire, r winding , increases with the temperature. the value of the winding resistance used should be at the operating temperature. )) t t ( 0042 . 0 1 ( r p c 20 h ) c 20 ( winding ) ht ( winding ? + = eq. 8 where: t h = temperature of wire under full load t 20c = ambient temperature r wind ing(20c) = room temperature winding resistance (usually specified by the manufacturer) output capacitor selection the type of the output capacitor is usually determined by its equivalent series resistance ( e sr ). voltage and rms current capability are two other important factors for selecting the output capacitor. recommended capacitor types are ceramic, low - esr aluminum electrolytic, os - con and poscap. the output capacitor?s esr is usually the main cause of the output ripple. the output capacitor esr also affects the control loop from a stability point of view.
micrel, inc. mic2 4051 november 2012 18 m9999 -1 12612 -a the maximum value of esr is calculated: ) pp ( l ) pp ( out c i v esr out ? ? eq. 9 where: v out(pp) = peak - to - peak output voltage ripple ?, l(pp) = peak - to - peak inductor current ripple the total output ripp le is a combination of the esr and output capacitance. the total ripple is calculated in equation 10: ( ) 2 c ) pp ( l 2 sw out ) pp ( l ) pp ( out out esr i 8 f c i v ? + ? ? ? ? ? ? ? ? ? = ? eq. 10 where: d = duty cycle c out = output capacitance value f sw = switching frequency as described in the ?theory of opera tion? subsection in functional description , the mic24051 requires at least 20mv peak - to - peak ripple at the fb pin to make the g m amplifier and the error comparator behave properly. also, the output voltage ripple should be in phase with the inducto r current. therefore, the output voltage ripple caused by the output capacitors value should be much smaller than the ripple caused by t he output capacitor esr. if low - esr capacitors, such as ceramic capacitors, are selected as the output capacitors, a rip ple injection method should be applied to provide the enough feedback voltage ripple. please refer to the ?ripple injection? subsection for more details. the voltage rating of the capacitor should be twice the output voltage for a tantalum and 20% greater for aluminum electrolytic or os - con. the output capacitor rms current is calculated in equation 11 : 12 i i ) pp ( l ) rms ( c out ? = eq. 11 the power dissipated in the output capacitor is: out out out c ) rms ( c c ( diss esr i ) p = eq. 12 input capacitor selection the input c apacitor for the power stage input v in should be selected for ripple current rating and voltage rating. tantalum input capacitors may fail when subjected to high inrush currents, caused by turning the input supply on. a tantalum input capacitor?s voltage r ating should be at least two times the maximum input voltage to maximize reliability. aluminum electrolytic, os - con, and multilayer polymer film capacitors can handle the higher inrush currents without voltage de - rating. the input voltage ripple will prima rily depend on the input capacitor?s esr. the peak input current is equal to the peak inductor current, so: in c ) pk ( l in esr i v = ? eq. 13 the input capacitor must be rated for the input current ripple. the rms value of input capacitor current is deter mined at the maximum output current. assuming the peak - to - peak inductor current ripple is low: ) d 1 ( d i ) rms ( i ) max ( out c in ? eq. 14 the power dissipated in the input capacitor is: in in in c ) rms ( c ) c ( diss esr i p = eq. 15 ripple injection the v fb ripple required for p roper operation of the mic24051 g m amplifier and error comparator is 20mv to 100mv. however, the output voltage ripple is generally designed as 1% to 2% of the output voltage. for a low output voltage, such as a 1v, the output voltage ripple is onl y 10mv to 20mv, and the feedback voltage ripple is less than 20mv. if the feedback voltage ripple is so small that the g m amplifier and error comparator can?t sense it, then the mic24051 will lose control and the output voltage is not regulated. in order to have some amount of v fb ripple, a ripple injection method is applied for low output voltage ripple applications.
micrel, inc. mic2 4051 november 2012 19 m9999 -1 12612 -a the applications are divided into three situations according to the amount of the feedback voltage ripple: 1. enough ripple at the feed back voltage due to the large esr of the output capacitors. as shown in figure 5 , the converter is stable without any ripple injection. the feedback voltage ripple is: ) pp ( l c ) pp ( fb i esr 2 r 1 r 2 r v out ? + = ? eq. 16 where i l(pp) is the peak - to - peak value of the induct or current ripple. 2. inadequate ripple at the feedback voltage due to the small esr of the output capacitors. the output voltage ripple is fed into the fb pin through a feedforward capacitor c ff in this situation, as shown in figure 6 . the typical c ff value is between 1nf and 100nf. with the feedforward capacitor, the feedback voltage ripple is very close to the output voltage ripple: ) pp ( l ) pp ( fb i esr v ? ? eq. 17 3. virtually no ripple at the fb pin voltage due to the very - low esr of the output capacitors. figure 5 . enough ripple at fb figure 6 . inadequate ripple at fb figure 7 . invisible ripple at fb in this situation, the output voltage ripple is less than 20mv. therefore, additional ripple is injected into the fb pin from the switching no de sw via a resistor r inj and a capacitor c inj , as shown in figure 7 . the injected ripple is: ? = ? sw div in ) pp ( fb f 1 ) d 1 ( d k v v eq. 18 2 r // 1 r r 2 r / 1 r k inj div + = eq. 19 w here : v in = power stage input voltage d = duty cycle f sw = switching frequency 2 = (r1//r2//r inj ) c ff in equations 18 and 19 , it is assumed that the time constant associated with c ff must be much greater than the switching period: 1 t fsw 1 << = eq. 20 if the voltage divider resistors r1 and r2 are in the k range, a c ff of 1nf to 100nf can easily satisfy the large time constant requirements. also, a 100nf injection capacitor c inj is used in order to be considered as short for a wide range of the frequencies.
micrel, inc. mic2 4051 november 2012 20 m9999 -1 12612 -a the process of sizing the ripple injection resistor and c apacitors is: step 1. select c ff to feed all output ripples into the feedback pin and make sure the large time constant assumption is satisfied. typical choice of c ff is 1nf to 100nf if r1 and r2 are in k range. step 2. select r inj according to the expected feedback voltage ripple using equation 19 . ) d 1 ( d f v v k sw in ) pp ( fb div ? ? = eq. 21 then the value of r inj is obtained as: ? ? ? ? ? ? ? ? ? = 1 1 ) 2 // 1 ( div k r r rinj eq. 22 step 3. sel ect c inj as 100nf, which could be considered as short for a wide range of the frequencies. setting output voltage the mic24 051 requires two resistors to set the out put voltage as shown in figure 8 . the output voltage is determined by equation 23: ? ? ? ? ? ? + = 2 r 1 r 1 v v fb out eq. 23 where v fb = 0.8v. a typical va lue of r1 can be between 3k and 10k. if r1 is too large, it may allow noise to be introduced into the voltage feedback loop. if r1 is too small, it will decrease the efficiency of the power supply, especially at light loads. once r1 is selected, r2 can b e calculated using: fb out fb v v 1 r v 2 r ? = eq. 24 figure 8 . voltage - divider configuration in addition to the external ripple injection added at the fb pin, internal ripple injection is added at the inverting input of the comparator inside the mi c24 051 , as shown in figure 9 . the inverting input voltage v inj is clamped to 1.2v. as v out is increased, the swing of v inj will be clamped. the clamped v inj reduces the line regulation because it is reflected as a dc error on the fb terminal. therefore, th e maximum output voltage of the mic24051 should be limited to 5.5v to avoid this problem. figure 9 . internal ripple injection thermal measurements measuring the ic?s case temperature is recommended to insure it is within its operating limits. although this might seem like a very elementary task, it is easy to get erroneous results. the most common mistake is to use the standard thermal couple that comes with a thermal meter. this thermal couple wire gauge is large, typically 22 gauge, and beha ves like a heatsink, resulting in a lower case measurement.
micrel, inc. mic2 4051 november 2012 21 m9999 -1 12612 -a two methods of temperature measurement are using a smaller thermal couple wire or an infrared thermometer. if a thermal couple wire is used, it must be constructed of 36 gauge wire or higher then (smaller wire size) to minimize the wire heat - sinking effect. in addition, the thermal couple tip must be covered in either thermal grease or thermal glue to make sure that the thermal couple junction is making good contact with the case of the ic. omega brand thermal couple (5sc - tt - k - 36- 36) is adequate for most applications. wherever possible, an infrared thermometer is recommended. the measurement spot size of most infrared thermometers is too large for an accurate reading on a small form factor ics. ho wever, an ir thermometer from optris has a 1mm spot size, which makes it a good choice for measuring the hottest point on the case. an optional stand makes it easy to hold the beam on the ic for long periods of time.
micrel, inc. mic2 4051 november 2012 22 m9999 -1 12612 -a pcb layout guidelines warning!!! to mi nimize emi and output noise, follow these layout recommendations. pcb layout is critical to achieve reliable, stable and efficient performance. a ground plane is required to control emi and minimize the inductance in power, signal and return paths. the fol lowing guidelines should be followed to insure proper operation of the mic24051 regulator. ic ? a 2.2f ceramic capacitor, which is connected to the pv dd pin, must be located right at the ic. the pvdd pin is very noise sensitive and placement of the capacitor is very critical. use wide traces to connect to the pv dd and pgnd pins. ? a 1 f ceramic capacitor must be placed right between vdd and the signal ground sgnd. the sgnd must be connected directly to the ground planes. do not route the sgnd pin to th e pgnd pad on the top layer. ? place the ic close to the point - of - load (pol). ? use fat traces to route the input and output power lines. ? signal and power grounds should be kept separate and connected at only one location. input capacitor ? place the input capa citor next. ? place the input capacitors on the same side of the board and as close to the ic as possible. ? keep both the pvin pin and pgnd connections short. ? place several vias to the ground plane close to the input capacitor ground terminal. ? use either x7r or x5r dielectric input capacitors. do not use y5v or z5u type capacitors. ? do not replace the ceramic input capacitor with any other type of capacitor. any type of capacitor can be placed in parallel with the input capacitor. ? if a tantalum input capacitor is placed in parallel with the input capacitor, it must be recommended for switching regulator applications and the operating voltage must be derated by 50%. ? in ?hot - plug? applications, a tantalum or electrolytic bypass capacitor must be used to limit the over - voltage spike seen on the input supply with power is suddenly applied. inductor ? keep the inductor connection to the switch node (sw) short. ? do not route any digital lines underneath or close to the inductor. ? keep the switch node (sw) away from the feedback (fb) pin. ? the cs pin should be connected directly to the sw pin to accurate sense the voltage across the low - side mosfet. ? to minimize noise, place a ground plane underneath the inductor. ? the inductor can be placed on the opposite side of the pcb w ith respect to the ic. it does not matter whether the ic or inductor is on the top or bottom as long as there is enough air flow to keep the power components within their temperature limits. the input and output capacitors must be placed on the same side o f the board as the ic. output capacitor ? use a wide trace to connect the output capacitor ground terminal to the input capacitor ground terminal. ? phase margin will change as the output capacitor value and esr changes. contact the factory if the output capac itor is different from what is shown in the bom. ? the feedback trace should be separate from the power trace and connected as close as possible to the output capacitor. sensing a long high - current load trace can degrade the dc load regulation. optional rc s nubber ? place the rc snubber on either side of the board and as close to the sw pin as possible.
micrel, inc. mic2 4051 november 2012 23 m9999 -1 12612 -a evaluation board schematic figure 10 . schematic of mic24051 evaluation board (j11 , r13, r15 are for testing purposes)
micrel, inc. mic2 4051 november 2012 24 m9999 -1 12612 -a evaluation board schematic ( continued) figure 11 . schematic of mic24051 evaluation board (optimized for smallest footprint)
micrel, inc. mic2 4051 november 2012 25 m9999 -1 12612 -a bill of materials item part number manufacturer description qty. c1 open c2 , c3 12103c475kat2a avx ( 1 ) 4.7f ceramic capacitor, x 7r, size 12 10, 25 v 2 grm32dr71e475ka61k murata ( 2 ) c3225x7r1e475k tdk ( 3 ) c5, c13, c15 open c4 12106d107mat2a avx 100 f ceramic capacitor, x5r, size 1210, 6.3v 1 grm32er60j107me20l murata c3225x5r0j107m tdk c6, c7, c10 06035c104kat2a avx 0.1f c eramic capacitor, x7r, size 0603, 50v 3 grm188r71h104ka93d murata c1608x7r1h104k tdk c8 0603zc10 5 kat2a avx 1.0f ceramic capacitor, x7r, size 0603 , 10v 1 grm188r71a105ka61d murata c1608x7r1a105k tdk c9 0603zd225k at2a avx 2.2f ceramic cap acitor, x5r, size 0603 , 10v 1 grm188r61a225ke34d murata c1608 x 5 r1a225k tdk c12 06035c472 kaz2a avx 4.7 nf ceramic capacitor, x7r, size 0603, 50v 1 grm188r71h472 k murata c1608x7r1h472 k tdk c14 b41851f7227m epcos ( 4 ) 220 f aluminum capacitor, 35 v 1 c11, c16 open d1 sd103 a ws mcc (5) 40v, 350ma, schottky diode, sod323 1 sd103 a ws -7 d iodes i nc ( 6 ) sd103 a ws vishay ( 7 ) l1 hcf1305 - 2r2 - r cooper bussmann ( 8 ) 2.2 h inductor, 15 a saturation current 1 r1 crcw06032r21fkea vishay dale 2.21 ? resistor, size 0603, 1% 1 r2 crcw0 6032 r 00 fkea vishay dale 2.00 ? resistor , size 0 603 , 1% 1 r3 crcw060319k6 fkea vishay dale 19.6 k ? resistor, size 0603, 1% 1 notes: 1. avx: www.avx.com . 2. murata: www.murata.com . 3. tdk: www.tdk.com . 4. epcos: www.epcos.com . 5. mcc : www.mccsemi.com . 6. diode inc.: www .diodes.com . 7. vishay: www.vishay.com . 8. cooper bussmann : www.cooperbussmann.com .
micrel, inc. mic2 4051 november 2012 26 m9999 -1 12612 -a bill of materials (continued) item part number manufacturer description qty. r4 crcw06032k49 fkea vishay dale 2.49 k ? resistor, size 0603, 1% 1 r 5 crcw0603 20k0 fkea vishay dal e 20.0 k ? resistor, size 0603, 1% 1 r 6, r14, r17 crcw 060310k0 fkea vishay dale 10.0k ? resistor, size 0603, 1% 3 r 7 crcw0603 4k99 fkea vishay dale 4.99 k ? resistor, size 0603, 1% 1 r 8 crcw06032k87 fkea vishay dale 2.87 k ? resistor, size 0603, 1% 1 r 9 crcw 06032 k006 fkea vishay dale 2.00k ? resistor, size 0603, 1% 1 r 10 crcw0603 1k18 fkea vishay dale 1.18 k ? resistor, size 0603, 1% 1 r 11 crcw0603 806r fkea vishay dale 806? resistor, size 0603, 1% 1 r 12 crcw 0603475r fkea vishay dale 475 ? resistor, size 0603, 1% 1 r 13 crcw06030000 fk ea vishay dale 0 ? resistor, size 0603, 5 % 1 r 15 crcw0603 49r9 fkea vishay dale 49.9 ? resistor, size 0603, 1% 1 r16, r18 crcw0 603 1r21fkea vishay dale 1.21 ? resistor , size 0 603 , 1% 2 r20 open all reference designators ending with ?a? open u1 mic24051 yjl micrel. inc. ( 9 ) 12v, 6a high - efficiency buck regulator 1 note : 9. micrel, inc.: www.micrel.com .
micrel, inc. mic2 4051 november 2012 27 m9999 -1 12612 -a pcb layout rec ommendations figure 12 . mic24051 evaluation board top layer figure 13 . mic24051 evaluation board mid - layer 1 (ground plane)
micrel, inc. mic2 4051 november 2012 28 m9999 -1 12612 -a pcb layout recommendations (continued) figure 14 . mic24051 evaluation board mid - layer 2 figure 15 . mic24051 evaluation board bottom layer
micrel, inc. mic2 4051 november 2012 29 m9999 -1 12612 -a package information (1) 28- pin 5mm x 6mm qfn ( jl) note: 1. package information is correct as of the publication date. for updates and most current information, go to www.micrel.com .
micrel, inc. mic2 4051 november 2012 30 m9999 -1 12612 -a micrel, inc. 2180 fortune drive san jose, ca 95131 usa tel +1 (408) 944 - 0800 fax +1 (408) 474 - 1000 web http://www.micrel .com micrel makes no representations or warranties with respect to the accuracy or completeness of the information furnished in th is data sheet. this information is not intended as a warranty and micrel does not assume responsibility for its use. micre l reserves the right to change circuitry, specifications and descriptions at any time without notice. no license, whether express, implied, arising by estoppel or otherwise, to any intellectual property rights is granted by this document. except as provi ded in micrel?s terms and conditions of sale for such products, micrel assumes no liability whatsoever, and micrel disclaims any express or implied warranty relating to the sale and/or use of micrel products including liability or warranties relating to fi tness for a particular purpose, merchantability, or infringement of any patent, copyright or other intellectual property righ t . micrel products are not designed or authorized for use as components in life support appliances, devices or systems where mal function of a product can reasonably be expected to result in personal injury. life support devices or systems are devices or syste ms that (a) are intended for surgical implant into the body or (b) support or sustain life, and whose failure to perform can be reasonably expected to result in a signific ant injury to the user. a purchaser?s use or sale of micrel products for use in life support appliances, devices or systems is a purchaser?s own risk and purchaser agrees to fully indemnify micrel for any damages resulting from such use or sale. ? 2012 micrel, incorporated.

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